Low noise charge pump method and apparatus

ABSTRACT

A charge pump method and apparatus is described having various aspects. Noise injection from a charge pump to other circuits may be reduced by limiting both positive and negative clock transition rates, as well as by limiting drive currents within clock generator driver circuits, and also by increasing a control node AC impedance of certain transfer capacitor coupling switches. A single-phase clock may be used to control as many as all active switches within a charge pump, and capacitive coupling may simplify biasing and timing for clock signals controlling transfer capacitor coupling switches. Any combination of such aspects of the method or apparatus may be employed to quiet and/or simplify charge pump designs over a wide range of charge pump architectures.

CROSS REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of 35 USC §120 as a divisionalapplication of co-pending U.S. application Ser. No. 10/658,154, “LowNoise Charge Pump Method and Apparatus” filed Sep. 8, 2003, and thecontents of this application to which priority is claimed is herebyincorporated in its entirety by reference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention generally relates to electronic power supplies, and morespecifically to capacitive energy transfer DC-to-DC converters (DC/DCconverters), such as charge pumps.

2. Related Art

DC/DC converter power supply circuits provide a DC output voltage basedupon a DC source voltage. The output is typically at a different voltagethan the input. As the term is used herein, DC/DC converters do notencompass voltage reduction regulator circuits that use a linear passdevice, but rather involve energy transfer from input to output throughan energy storage device, such as a capacitor or an inductor.

The DC/DC converters of interest herein are charge pumps, which obtainenergy for the output voltage primarily by means of capacitive transferfrom the source to the output. An inductor is not generally the primaryenergy transfer device in a charge pump, though of course hybrid devicesare possible that employ inductive energy transfer in addition tocapacitive energy transfer. A charge pump may derive an output voltagethat is higher than a source voltage, or that is inverted from a sourcevoltage, or that is referenced to a different voltage than the sourcevoltage, and indeed may do all of these things concurrently.

Charge pumps may be implemented for a wide variety of purposes. They arewell suited for integrated circuit fabrication because the devices andelements required are compatible with most integrated circuitfabrication techniques. For example, a charge pump may be employed togenerate a negative gate bias supply for an integrated circuit thatswitches an antenna between send and receive circuitry of a transceiver,as shown in FIG. 1. Many wireless transceivers, such as cellulartelephones, employ a single antenna for both receiving and transmitting.While such systems are receiving, an antenna 102 must be coupled toreceive circuitry that may include, for example, a filter 104 and a lownoise amplifier 106, to provide the received signal for furtherprocessing. However, while such systems are transmitting, the antenna102 must be disconnected from the sensitive receive circuitry andcoupled instead to relatively high power transmit circuitry. Thetransmit circuitry may include, for example, a power amplifier 108 and atransmit filter 110 to process a transmit signal.

A RF switch 112 may be used to perform such antenna switching functions.Ideally, such switches may be integrated together with the receiveand/or transmit circuitry, and in any event are desirably very small,due to space limitations in portable transceivers such as mobiletelephones and handy talkies. In order to achieve good performance fromswitching devices, such as MOSFETs, used to implement such RF switches,many designs need a special bias supply that extends negatively belowthe supply rails of the transmit and receive circuitry, such as a −3Vsupply. In view of the space and cost constraints of transceiver unitssuch as mobile telephones, a charge pump is particularly suitable forgenerating such a bias supply, because it can be readily integrated intoa very small circuit.

The RF switch 112 conveys relatively high power signals to the antenna102 during transmission. However, during receive, the signal passed bythe RF switch 112 may be measured in tens of nanovolts. Sharp noisetransitions may have an extremely broad Fourier frequency content, andthus even signals at amplitudes on the order of millivolts may interfereunacceptably with reception if the signals have extremely fast edges.While the filter 104 can remove some noise, it is important that the RFswitch 112 not introduce noise, particularly noise having componentsnear the center frequency of the received signal. Thus, thereceive/transmit switch of FIG. 1 illustrates one of many circumstancesin which a charge pump may be desired for a circuit that nonethelessrequires extremely low noise.

Unfortunately, noise generation is one of the most common drawbacks ofcharge pumps. Current spikes are typically coupled into both input andoutput supplies, together with voltage ripples and spikes. When a chargepump is integrated together with other devices, such electronic noisemay be coupled throughout the circuitry of the integrated device by avariety of mechanisms that are difficult to control. Thus, a need existsfor charge pumps that avoid generating excessive noise, so as to reducecharge pump noise injection into source supplies, output supplies, andrelated circuits.

The method and apparatus presented herein address this need for alow-noise charge pump. Various aspects of the method and apparatusdescribed herein will be seen to provide further advantages, as well,for the design and construction of charge pumps that are relatively freeof noise spurs.

SUMMARY

A charge pump method and apparatus for DC-to-DC conversion is set forthby which an output is generated by alternately coupling a transfercapacitor to an input supply and then to the output. A charge pump clockoutput is generated to control a transfer capacitor coupling switch in acharge pump circuit. The charge pump method (or apparatus) furtherincludes a combination of one or more aspects of the method orapparatus, as set forth below, in order to reduce noise or otherwiseimprove the design.

One aspect of the charge pump method described herein includesgenerating an output supply by transferring charge from a source voltageto a transfer capacitor (“TC”) alternately with transferring charge fromthe TC to the output supply. A TC-coupling switch (“TCCS”) circuit is aswitch that couples the TC to a supply under control of a charge pumpclock (“CPClk”). This method includes coupling the TC to the outputsupply during discharge periods via a discharging TCCS circuit, andactively limiting a rate of voltage change of the CPClk output duringboth positive and negative transitions.

An aspect of the charge pump apparatus described herein may be employedto generate an output voltage supply within a circuit. This aspectincludes a transfer capacitor (“TC”) and a plurality of TC couplingswitches (“TCCSs”), each TCCS controlled by a charge pump clock(“CPClk”) output. The apparatus also has a CPClk generation circuit,which includes circuitry that limits a rate of rise of the CPClk outputand circuitry that limits a rate of fall of the CPClk output. The TCCSsare coupled to the TC, and are controlled to couple the TC to a voltagesource during periodic first periods, and to couple the TC to the outputvoltage during periodic second periods that are not concurrent with thefirst periods.

Another aspect of the charge pump apparatus described herein may be alsobe used for generating an output voltage supply within a circuit. Thisaspect includes a transfer capacitor (“TC”) coupled alternately betweensource connections and output connections, and a plurality of activeswitches that are each switchable under control of a charge pump clock(“CPClk”) output to conduct, or not. This aspect includes a CPClkgenerating circuit that has an active driver circuit configured tosource current to, and sink current from, a driver output node, togetherwith circuitry to limit the source current, and circuitry to limit thesink current, of the active driver circuit.

Another aspect of the charge pump described herein is a method ofalternately transferring charge from a source voltage to a transfercapacitor (“TC”), and from the TC to an output supply, to generate anoutput supply. The method includes coupling the TC to the output supplyvia a discharging switch under control of a charge pump clock (“CPClk”)output. The method also includes limiting source and sink currentsbetween a driver circuit and a corresponding output node, within agenerator circuit of the CPClk.

Yet another aspect of the charge pump method described hereinalternately transfers charge from a source voltage to a transfercapacitor (“TC”), and from the TC to an output supply. This methodincludes coupling the TC to the output supply during discharge periodsvia a discharging switch circuit under control of a single phase chargepump clock (“CPClk”) output, and coupling the TC to the voltage sourcevia a charging switch circuit under control of the single phase CPClkoutput.

A related aspect of the charge pump apparatus described herein may beused for generating an output voltage supply. This apparatus includes atransfer capacitor (“TC”), one or more source switching devices inseries between the TC and the source voltage, and one or more outputswitching devices in series between the TC and the output supply. Asingle phase output of a charge pump clock (“CPClk”) generating circuitis coupled to all of the source switching devices to cause them toconduct only during charge periods, and is also coupled to all of theoutput switching devices to cause them to conduct only during dischargeperiods, wherein the charge and discharge periods alternate and do notoverlap.

A further aspect of the charge pump method described herein may generatean output supply by alternately transferring charge from a sourcevoltage to a transfer capacitor (“TC”), and from the TC to the outputsupply. This method includes coupling a first charge pump clock(“CPClk”) output to a TC charging switch control node via a firstcapacitive coupling network, and coupling the TC to the source voltageduring charge periods under control of the first CPClk output. Themethod also includes coupling a second CPClk output to a TC dischargingswitch control node via a second capacitive coupling network, andcoupling the TC to the output supply during discharge periods thatnonconcurrently alternate with the charge periods, under control of thesecond CPClk output.

A further aspect of the charge pump apparatus described herein may beused for generating an output voltage supply, and includes a transfercapacitor (“TC”). This apparatus includes one or more source switchingdevices, having a corresponding control node, disposed in series betweenthe TC and a source supply, and also includes one or more outputswitching devices, having corresponding control nodes, disposed inseries between the TC and the output supply. This apparatus alsoincludes a capacitive coupling circuit coupling a charge pump clockoutput to the control node of a source switching device or to thecontrol node of an output switching device.

Yet a further method aspect of the charge pump described herein includessteps of disposing an output hot switch between a transfer capacitor(“TC”) node and a charge pump output, and disposing an output commonswitch between the opposite TC node and a common connection of thecharge pump output. This method also includes establishing a highercontrol node AC impedance for the hot switch, as compared to a controlnode AC impedance for the output common switch. A related aspectprovides apparatus generally corresponding to this method.

Embodiments of the charge pump method or apparatus may employ anycombination of individual aspects of the method or apparatus, and may beconfigured in a wide range of charge pump architectures andconfigurations.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will be more readily understood byreference to the following figures, in which like reference numbers anddesignations indicate like elements.

FIG. 1 is a block diagram illustrating use of a RF switch.

FIG. 2 is a block diagram illustrating basic charge pump operations.

FIG. 3 is a charge pump schematic diagram illustrating some of thecurrent paths in which noise may be reduced by employing aspects of themethod and apparatus described herein.

FIG. 4 is a schematic diagram of exemplary current-limited inverters.

FIG. 5 is a schematic diagram of a current-starved oscillator for acharge pump.

FIG. 6 is a schematic diagram of a charge pump in which the activeswitches are coupled to control signals by means of capacitive coupling.

FIG. 7 is a schematic diagram of a charge pump employing passiveswitches.

FIG. 8 is a schematic diagram of charge pump switches having a controlnode capacitively coupled to a charge pump clock output.

FIG. 9 is a block diagram illustrating alternative charge pumpconfigurations.

FIG. 10 is a block schematic diagram illustrating techniques for usingcharge pumps to produce arbitrary output voltages.

DETAILED DESCRIPTION Overview—Charge Pump Configurations

FIG. 2 illustrates some basic charge pump operations. Filteringcapacitors, typically present on each input and output supply, areomitted from FIG. 2 to avoid confusion with an energy transfer capacitorC_(T) 202. When S1 204 and S2 206 are closed, C_(T) 202 is connected tothe source DC supply connections V_(S)+ 208 and V_(S)− 210, and thusC_(T) 202 is charged to the voltage V_(S), having a value [(Vs+)−(Vs−)].S1 204 and S2 206 are then opened, permitting C_(T) 202 to float. Next,S3 212 and S4 214 are closed, causing the floating transfer capacitorC_(T) 202 to “fly” to the output supply (for which reason suchcapacitors are also called fly capacitors). The charged C_(T) is thusconnected to output supply connections Vo1+ 216 and Vo1− 218 (outputVo1). Presuming that the voltage of output Vo1 is smaller than thevoltage stored on C_(T) 202, C_(T) 202 discharges, transferring energyto Vo1, which may be an output supply.

Many different output configurations are possible using just these fourswitches S1 204, S2 206, S3 212 and S4 214, together with the transfercapacitor C_(T) 202. A first example is a voltage doublingconfiguration, wherein Vo1− 218 is connected to V_(S)+ 208, causing Vo1+216 to achieve a voltage roughly twice that of V_(S)+ 208 (both withrespect to V_(S)− 210, neglecting losses). A second example is a voltageinverting configuration, in which Vo1+ 216 is connected to V_(S)− 210,such that Vo1− will reach approximately −V_(S) (with respect to V_(S)−210). A third example is a supply isolating configuration, in whichneither Vo1+ 216 nor Vo1− 218 is tied through a fixed voltage connectionto a source voltage connection. Rather, the input lines may be isolatedfrom the output insofar as is permitted by the isolation capability ofthe switches S1 204 and S3 212, and of the switches S2 206 and S4 214.

The skilled person will understand certain features without a need forexplicit details. For example, because the transfer capacitor C_(T) 202must be disconnected from the output while connecting to the input,maintaining a reasonably constant voltage on the output generallyrequires a storage device. Such storage device typically comprises afilter capacitor, which is not shown in FIG. 2. As another example,current and voltage for the output may vary depending upon many factors.The skilled person may make allowance for such factors to anticipate avoltage of the output, or may choose to regulate the voltage of theoutput. Such regulation is not shown, but may, for example, comprisecontrolling the frequency of “pump” cycles during which C_(T) 202 isfirst charged to V_(S) and then discharged to voltage Vo1. Regulationmay also comprise controlling a value of the voltage source for thecharge pump, as described with respect to FIG. 10 below.

Additional output connection switches, such as S5 220 and S6 222, mayenable a charge pump to provide other voltages, multiplexing the use ofa single transfer or fly capacitor C_(T) 202. For example, charge anddischarge cycles may alternate without overlapping. C_(T) may be chargedto V_(S) during each charge cycle. However, during one discharge cycle,C_(T) may be discharged to Vo2 (Vo2+ 224 to Vo2− 226), while duringanother discharge cycle C_(T) is discharged to Vo1. In this manner,three voltages of roughly equal value may be created—V_(S), Vo1 and Vo2.The connections made between these voltages determines whether thecharge pump circuit functions as a voltage tripler, or as adouble-voltage inverter, an isolated 2*V_(S) supply, two isolatedoutputs, etc. These principles may be extended to further outputcombinations.

In typical implementations, each of the switches S1 204, S2 206, S3 212and S4 214 (as well as other switches if used, such as S5 220 and S6222) may comprise an appropriate transistor, such as a MOSFET. However,in many circumstances it is possible to substitute a simple diode for aswitch, when the voltage and current flow requirements of the particularconfiguration of a specific charge pump circuit permit.

The basic charge pump architecture illustrated in FIG. 2 may also bereplicated for operation in a “push-pull” fashion. For this purpose, asecond C_(T) may be connected to V_(S) while the first C_(T) isconnected to Vo1, and then such second C_(T) may be discharged to Vo 1the first C_(T) is connected to V_(S). Such techniques may provide moreconstant current capacity to the output supply, and may aid in reducingoutput voltage ripple. Thus, charge pump DC/DC converters may bedesigned in a wide range of configurations to provide many differentoutputs from a single DC supply.

Charge Pump Noise

As noted above, noise is one of the most common problems associated withcharge pumps. FIG. 3 illustrates an exemplary charge pump architecture,showing some of the paths in which noise currents may typically occur.The charge pump of FIG. 3 includes a transfer capacitor C_(T) 202,together with source supply connections Vs+ 208 and Vs− 210, which aresubstantially as shown in FIG. 2, and accordingly are identicallynumbered. FIG. 3 also includes output supply connections Vo+ 316 and Vo−318. The fly capacitor coupling switches, which couple C_(T) 202 to thesource supply Vs or to the output supply Vo, are shown as MOSFET devicescontrolled by a signal applied to a control node (the device gate). Thefly capacitor coupling switches include a P-channel switch 304 undercontrol of a first clock output 354, a P-channel switch 306 undercontrol of a second clock output 356, an N-channel switch 312 undercontrol of a third clock output 362, and an N-channel switch 314 undercontrol of a fourth clock output 364. The four charge pump clock outputs(354, 356, 362, and 364) are generated in a charge pump clock generatorcircuit 350. A filter capacitor 330 for the output supply Vo, as well asa current load 332 on Vo, are also shown.

Four ammeters are represented to illustrate paths in which noisecurrents may occur. From these paths, noise may be unintentionallycoupled other circuits associated with the charge pump. For example,voltage noise may be coupled into other circuits through common supplyconnections, or via parasitic capacitive coupling, while current noisemay be injected into other circuits through inductive coupling, or via ashared impedance. A source loading ammeter 342 is disposed between thesource supply Vs+ 208 and the switch 304, and an output charging ammeter344 is disposed between the switch 312 and the output filter capacitor330. Clock generator ammeters are disposed between the clock generator350 and its supply source, with ammeter 346 between Vs+ 208 and clockgenerator 350, and ammeter 348 between the clock generator 350 and Vs−210. The explanation of charge pump noise currents, which is set forthbelow with reference to these four ammeters, is representative innature, and is not intended to be comprehensive.

In FIG. 3, a charging period begins at the moment when both the switches304 and 306 are closed, such that the C_(T) 202 is connected across thesource supply and the ammeter 342 registers the charging current forC_(T) 202. An amplitude of a spike in the charging current is passivelylimited by the source impedance of the supply Vs and the parasiticresistance of C_(T) 202 itself, as well as by the difference between Vsand the voltage on the C_(T) 202. The spike amplitude may also belimited by active control of conduction by the switches 304 and 306. Inaddition to controlling peak amplitude, a rate of rise of the chargingcurrent, di/dt, may be limited by limiting the speed with which theswitches 304 and 306 are turned on.

At the end of the charging period, the switches 304 and 306 will beturned off. A change in current will register in the ammeter 342 ifC_(T) 202 is not fully charged at this time, with a di/dt slope thatdepends in part upon the speed with which the series combination ofswitches 304 and 306 is turned off. Because the first clock output 354and the second clock output 356 are coupled to corresponding controlnodes of switches 304 and 306, respectively, the outputs 354 and 356will be driven up to a voltage greater than their threshold voltages,which will induce a nonconducting off state in these switches. To driveup the voltage of the control nodes, these clock outputs source currentto drive the gate capacitance (and/or other parasitic capacitances andimpedances) of the corresponding switch. The drive current will registerin the ammeter 346. Due to the parasitic capacitances, a high dv/dt gatedrive will typically cause a significant capacitive current spike in thecorresponding ammeter 346.

After the switches 304 and 306 are off, a discharge period may ensuethat does not overlap the charge period. To begin the discharge period,the third clock 362 and the fourth clock 364 will raise voltages ofcontrol nodes of the switches 314 and 312, respectively, above a Vgsthreshold voltage to turn each switch on to a conducting state. Theresulting capacitive gate current of the switches 312 and 314 willregister in the ammeter 346, with a peak value and di/dt that depend inpart upon dv/dt of the clock output. In order to minimize powerdissipation in the switches, the clock edges in such circuits havetypically been made rather fast.

As the later of switches 312 and 314 turns on, C_(T) 202 beginsdischarging into Co 330 with a current that will register in the ammeter344. At the end of the discharge clock period, the third clock output362 and the fourth clock output 364 sink current to drive down to an offvoltage, causing gate currents, to discharge the gate capacitance of thecorresponding switches, that will register in ammeter 348. Finally,C_(T) 202 is disconnected from Co 330, which may cause another currentstep to register in the ammeter 344 if discharging of C_(T) 202 has notbeen completed.

All switches remain off until the clock outputs 354 and 356 sink currentfrom the gates of switches 304 and 306, respectively, to reduce thecontrol node voltage of those switches to a level that is negativelygreater than their Vgs thresholds, thereby causing the switches to turnon and begin the charging period. The sink current provided by the clockoutputs 354 and 356 to effect this change will register in the ammeter348. A di/dt of such current depends substantially upon a magnitude ofthe dv/dt at which these clock outputs transition negatively.

Each high di/dt current spike described above is likely to be coupledinto nearby circuitry, whether through mutual inductances, commonimpedances, or common connections. Reducing some or all of theidentified noise sources may be desirable for circuits, such as thatdescribed with respect to FIG. 1, which need a charge pump DC/DCconverter but also require very low noise. Various features of thefollowing figure, taken in combinations, may be employed to constructsuch a desirable charge pump. Other benefits will also become apparent,such as the convenience and simplicity of control by a single-phaseclock.

Quiet Inverting Charge Pump

FIGS. 4, 5 and 6 illustrate an exemplary quiet, inverting embodiment ofa charge pump. FIG. 4 illustrates exemplary circuitry for establishingcurrent-limited drive circuits (here, inverters), which may be usedwithin a charge pump clock generating circuit. FIG. 5 illustrates anexemplary charge pump clock generating circuit that utilizescurrent-limited, inverting drive circuits configured to form acurrent-starved ring oscillator. FIG. 6 illustrates using a singlecharge pump clock output to drive all of the active fly capacitorcoupling switches in an exemplary charge pump. Throughout thedescriptions that are set forth below, size references for someexemplary FET devices are indicated parenthetically as a channel width(W) and length (L), in micron units. These size references areappropriate for a low voltage silicon-on-insulator (“SOP') integratedcircuit, and are useful for relative size considerations. In anexemplary embodiment, threshold voltages for P-channel MOSFETs are −0.25V, and for N-channel MOSFETs are +0.5 V. Device sizes, thresholdvoltages, and other considerations may, of course, be adjusted asappropriate for varying designs and processes, and indeed alternativesto MOSFETS, such as BJTs or other amplifying/switching devices, may beused with appropriate adaptation.

FIG. 4 is a schematic diagram of a current limited, inverting drivecircuit. An input voltage supply is connected via Vin+ 402 and commonconnection 404. A reference current, limited by a resistor 406, isestablished through diode-connected MOSFETs 408 (W 7.5, L 2) and 410 (W7.5, L 4) to create a P-gate reference voltage 412 and an N-gatereference voltage 414. The drive circuit 416 has an input node 418 andan output node 420, and is formed by connecting a P-channel FET 422 (W5, L 0.8) in series with an N-channel FET 424 (W 2, L 0.8), as shown.The drive circuit 416 drives current to the output node 420, sourcing orsinking current to raise or lower the voltage of the output node.Circuitry is provided that limits the current drive capability of thedrive circuit 416, as compared to an absence of such devices. “Absence”of a series device should be understood to mean replacement by a directconnection, while “absence” of a parallel device should be understood tomean merely omission of such device.

The current source capability of the driver circuit 416 is limitedthrough a source current limit circuit 426, which may, for example,comprise a P-channel FET 428 (W 1.5, L 2). The FET 428 may begate-coupled to the P-gate reference 412 as a current mirror that limitscurrent, for example, to 1 μA or less. The current sink capability ofthe driver circuit 416 may be similarly limited by a sink current limitcircuit 430, comprised for example of an N-channel FET 432 (W 1.5, L 4)coupled to the N-gate reference 414 as a current mirror (e.g. limited to1 μA). Establishing 1 μA current limits in FETs 428 and 432 may requireabout 5 μA through reference current FETs 408 and 410, due to the 1:5size ratio of the exemplary corresponding devices. Equal magnitudes forsource and sink current limits may reduce undesirable harmonicgeneration. Any other appropriate techniques for creating currentlimited drive circuits may be employed in the alternative, such as usinglow-conductivity FETs or resistors, etc., to restrain either or both ofthe source and sink drive capability of such drive circuit.

Additional circuitry, shown connected by phantom lines to the P-gate andN-gate references 412 and 414, represents optional additionalcurrent-limited inverter circuits that may be coupled to the samecurrent reference voltages 412 and 414.

FIG. 5 is a schematic diagram of a ring oscillator 500, which will bereferred to as a “current-starved” ring oscillator operating with anexemplary period of 1 _([)IS. Inverters 502, 504 and 506 may beequivalent to the inverter 416 as shown in FIG. 4. Each limits currentfrom a source supply connection Vin+ 402 by means of a source currentlimit 508, 510 and 512, respectively, which may be equivalent to thesource current limit 426 shown in FIG. 4. Each inverter also limitscurrent from a sink supply connection common reference 404 by means of asink current limit 514, 516 or 518, respectively, each of which may beequivalent to the sink current limit 430 of FIG. 4. Output transitionrates (dv/dt) for the inverting drive circuit 502 are actively limitedby a capacitor 520 (e.g., 90 ff) coupled to its output node, inconjunction with the “current starving” effect of the source and sinkcurrent limits 508 and 514, respectively. Similarly, dv/dt of the outputnode of the inverting drive circuit 504 is also actively limited by acapacitor 522 (e.g., 90 ff) coupled thereto, in conjunction with thecurrent drive limit devices 510 and 516 of the inverting drive circuit504.

Output transition dv/dt of CLK output 524 of the driver circuit 506 isactively limited by the source and sink current limits 512 and 518, inconjunction with the combined distributed capacitance of each drivendevice, which is represented by phantom capacitor 526. A discretecapacitor may, of course, be added if appropriate. The devices thatconstrain dv/dt on the input node of driver circuit 506 also limit dv/dtof the CLK output 524. Series connection of the three inverting drivercircuits 502, 504, and 506, creates a ring oscillator. Thecurrent-starved ring oscillator 500 is an example of a charge pump clockgenerating circuit that includes circuits to reduce switching speeds ofsubstantially all driver circuits within the clock generator. Thecurrent-starved ring oscillator 500 is also an exemplary circuit forproducing a charge pump clock output 524 for which dv/dt is activelylimited in both negative and positive transitions. Voltage of the CLKoutput 524 may oscillate substantially rail-to-rail (e.g., between 0 toVin+) with low dv/dt transitions, and may have a significantly sine-likeshape.

FIG. 6 is a schematic diagram of a charge pump that may employ a singlephase of a limited dv/dt clock, such as the output CLK 524 of the clockgenerator 500 shown in FIG. 5. In an exemplary embodiment, Vin+ is +3Vwith respect to ground (common reference), while Vo− is −3V. P-channelMOSFETs 602 and 604 (W 20, L 0.8) conduct during a charge phase,connecting a fly capacitor 606 (e.g., 10 pf) to the source supply viaconnections Vin+ 402 and common reference 404. At a later time, during adischarge period, the FETs 602 and 604 will be off (nonconducting), andN-channel FETs 608 (W 20, L 0.8) and 610 (W 5, L 0.8) will conduct,connecting the fly capacitor 606 between a common reference plus side,and an intermediate output stored on a storage capacitor 614.

During this discharge period, the N-channel FET 608 couples one terminalof the fly capacitor 606 to a common reference (404) connection of theoutput supply. Concurrently, the N-channel FET 610 couples the oppositeterminal of the fly capacitor 606 to the connection of the output supply(the junction of FET 610 and capacitor 614) that is opposite to thecommon reference connection of the output supply. The area of the device610 is made much smaller (e.g., half as large, or less—one fourth aslarge in the exemplary embodiment) than the area of the device 608. Thesmaller size helps to reduce noise injection into the output, forexample by minimizing coupling of control signals through the gatecapacitance of the FET 610. For a typical FET, the device area is simplythe length times the width. If other processing parameters are constant,a control node AC impedance (to conducting control signals into theswitch) will vary approximately inversely with such area. However,control signal AC coupling impedance may be increased in other ways, forexample by reducing the parasitic gate-body capacitance throughincreases in the dielectric thickness or reduction in the dielectricconstant. Thus, the device 610 may be configured in various ways to haveat least twice the control node AC impedance of the device 608.

The voltage on storage capacitor 614 is filtered, by means of a resistor616 and filter capacitor 618, to provide a filtered output supplyvoltage Vo− with respect to common reference 404. The exemplaryembodiment provides a quiet output but relatively low current capacity.The skilled person will readily adjust the current capacity of a circuitsuch as that of FIG. 6, for example by employing small (to zero) valuesfor the resistor 616, or by replacing resistor 616 with a primarilyinductive impedance. Larger devices may also be employed for any or allof FETs 602, 604, 608 and 610, as well as a larger fly capacitor 606.

Each of the FETs 602, 604, 608 and 610 is an example of an activelycontrollable transfer capacitor coupling switch (“TCCS”). First, each isdisposed in series to couple a node of the transfer capacitor to thesource supply (for charging switches) or to the output supply (fordischarging switches). Second, the conductivity of each is activelycontrollable, under control of a signal applied to a control node of theswitch (here, the gate of the FET). Such a control node has asignificant impedance to primary conducting connections (here, thesource and drain connections).

As shown in FIG. 6, the gate (control node) of each TCCS 602, 604, 608and 610 may be capacitively coupled to the appropriate clock signal bymeans of a corresponding coupling capacitor in conjunction with a biasresistor. Direct coupling could be employed by use of appropriate levelshifting and isolation considerations. For the FETs 602, 604 and 608,capacitive coupling is effected by means of capacitors (e.g., 1 pf) 618,620 and 622, respectively, together with resistors (e.g., 10 MΩ) 626,628 and 630, respectively.

In the case of a FET 610, the coupling is effected by two couplingcircuits connected in series to reduce voltage stresses. Thus, acapacitor 636 (e.g., 2 pf) couples the CLK 524 to an intermediate nodewhere it is biased by a resistor 638 (e.g., 10 MΩ) to an intermediateaverage voltage level (common reference voltage, in this case). Fromthis intermediate node the charge pump clock output signal is coupledvia a capacitor 624 (e.g., 2 pf) to the gate of the FET 610, and biasedby a resistor 632 (e.g., 10 MΩ) to the source voltage of the FET. Themodified capacitive coupling circuit for the FET 610 provides an exampleof a modified circuit that differs in some regard, but functionssubstantially equivalently, to another. Any circuit illustrated ordescribed herein may be replaced (as needed) by such a modified circuit,if substantially the same function is performed. Extended voltagecapacity circuits, such as series or cascode-coupled transistors or theseries capacitors described above, and increased current capacitycircuits, such as parallel combinations of transistors, are examples ofsuch equivalent circuits that may be employed, in place of anyillustrated circuit, to satisfy particular design goals.

In the exemplary circuit, N-channel switches such as FETs 608 and 610are switched on concurrently while P-channel switches such as FETs 602and 604 are off, and the converse is also true. The gate thresholdvoltage of N-channel switches is positive (e.g., conductive forVgs>0.5V), while the gate threshold voltage of P-channel switches isnegative (e.g., conductive for Vgs<−0.25V). The active limitation of thedv/dt of transitions of CLK 524 therefore creates a period of time whenall switches are off (i.e., the duration of the transition from [averageof CLK 524 −0.25] to [average of CLK 524 +0.5V]. Single-phase clockdrive of a charge pump may be effected using different types of switchesthat have correspondingly different control voltage thresholds. However,bias voltages may need to be adjusted accordingly to avoid simultaneousconduction of devices, such as FETs 602 and 608, that are disposed inseries across a supply. The circuits and devices of FIG. 6 provide aconvenient approach for driving all actively controllable TCCSs of acharge pump from a single clock phase (e.g., CLK 524), while ensuringthat conduction periods for series fly capacitor coupling switches donot inadvertently overlap.

Each capacitive gate drive circuit shown in FIG. 6 has a time constant τwhich may be expressed in terms of drive periods as R*C/(clock driveperiod). In the exemplary circuit of FIG. 6, τ is approximately 10periods, because the switch drive signal CLK 524 has a period of about 1μS. However, in many circumstances τ may be varied widely without undulyaffecting operation of the circuit. τ may be enlarged as much asdesired, though this may adversely affect device area. In somecircumstances, τ may also be reduced, and allowed to range for exampledown to 5 periods, to 2 periods, to 1 period, or even to 0.1 period orless without unduly impairing performance. Such reduction of τ mayreduce the amplitude of the gate drive applied to the correspondingMOSFET, particularly if it is accomplished by reducing the size of thecoupling capacitor (e.g., 618, 620, 622 or 624). This is due in part tocapacitive voltage division between the coupling capacitor and theMOSFET gate capacitance. Such voltage division may reduce the magnitudeof the control node voltage (Vgs), and hence may reduce conductivity ofthe switches when they are nominally enabled. Reducing τ may also reducethe proportion of time within each cycle during which the correspondingswitch is turned on, which may in turn increase ripple voltages and/orload current capacity. The wide range from which τ may otherwise bechosen may be limited by such engineering considerations.

Alternative Charge Pump Embodiments

Charge pumps are extremely versatile, and aspects of the charge pumpmethod and apparatus herein may be employed in virtually any charge pumpconfiguration. Some alternative techniques are set forth below, each ofwhich may be implemented using aspects of the method and apparatusdescribed herein.

FIG. 7 is a schematic diagram of a charge pump that generates a negativevoltage Vo− 710, and provides an example of using passive transfercapacitor coupling switches. A clock source CLK 524 is provided, whichmay be generated, for example, by the circuit of FIG. 5. A transfercapacitor 702 charges via the clock source and via a passive switch 704when CLK 524 is at a maximum value. When CLK 524 is at a minimum value,the transfer capacitor 702 discharges via the clock sink and a passiveswitch 706.

Passive switch 704 may be an N-channel enhancement MOSFET connecteddrain to gate and having a low threshold voltage of about 0.5V, or maybe a diode (anode to capacitor 702, cathode to ground), for example alow-voltage Schottky diode. Passive switch 706 may similarly be adiode-connected N-channel enhancement FET, or may be a discrete diodewith anode to output capacitor 708, cathode to transfer capacitor 702.The dv/dt control of the drive CLK 524 effects quiet operation. However,there is some loss of efficiency and voltage output due to the non-zeroforward conduction voltage of the devices 704 and 706. Thus, if CLK 524is 3V p-p (e.g., oscillates between +3V and common), then Vo− 710 may beonly about −2.5V, even at light loads.

Passive switches similar to devices 704 or 706 may be substituted inplace of one or more active switches in many designs described herein,simplifying design but typically reducing output voltage. For example,in FIG. 6, either or both of the active switches 604 and 610, togetherwith their associated drive coupling circuitry, may be replaced bypassive devices similar to devices 704 or 706. Passive switches arecontrolled by a charge pump clock output, but not via a control node.Rather, connections to primary conduction nodes of such switches aredriven, through circuit operation controlled by a charge pump clockoutput, to voltage levels at which the devices passively switch on oroff.

FIG. 8 is a schematic diagram of an active p-switch 800 and of an activen-switch 830. Each is driven by a clock CLK 802. The p-switch 800, when“on,” connects an external supply (e.g., V+ 804) to a fly capacitor fora charge pump, represented by Cp 806. The n-switch 830, when “on,”connects a charge pump fly capacitor (or transfer capacitor),represented by Cp 814, to a supply connection such as V− 816. Thep-switch 800 couples CLK 802 to the gate of a P-channel MOSFET 808 via acoupling capacitor 812 and a bias resistor 810. With the bias resistor810 connected to the source of the FET 808, it is preferable that theFET 808 has a small negative threshold voltage (e.g., −0.25V). For then-switch 830, the gate of an N-channel MOSFET 818 is coupled to CLK 802via a coupling capacitor 822, with a bias resistor 820 setting theaverage Vgs to zero volts. In this circumstance, it is preferred thatthe threshold voltage for the FET 818 has a small positive value (e.g.0.5V). Depending upon voltage levels, these switches 800 and 830 mayemploy series capacitive coupling circuits, and/or cascode or otherextended-capability circuit equivalents. The effective τ of eachcapacitive coupling circuit may be set in accordance with the operatingfrequency of a charge pump in which it is used, ranging from 10 clockperiods or more down to 0.1 clock periods or less, as described withrespect to FIG. 6. The τ of the capacitive coupling circuit within eachn-switch and p-switch operating within a single charge pump undercontrol of the same clock (e.g., CLK 524) may have substantially thesame value. P-switches 800 and n-switches 830 may have differentthresholds while operating from a common clock signal, provided that thebias voltage is set properly to ensure that a p-switch and an n-switchconnected in series across a supply are precluded from concurrentconduction.

FIG. 9 is a block schematic diagram that illustrates “stacking” chargepump circuits to obtain higher or lower output voltages. Each switch 800and 830 may be as illustrated in FIG. 8, or as discussed in the textdescribing FIG. 8. The circuit of FIG. 9 may be configured as either avoltage tripler, or a voltage quadrupler, as follows: Va 904 may beconnected to a positive source voltage

Vs+, and Vb 906 may be connected to a common reference voltage (ground).Vc 908 may also be coupled to Vs+. Operation of the n-switches and thep-switches connected to a transfer capacitor 902 will cause Vd 910 to bedriven to 2*Vs+. Ve 914 may be connected to Vd 910, and Vf 916 may beconnected to ground. With those connections, an output voltage Vh 920will be driven to approximately 3*Vs+ by means of fly capacitor 912 ifVg 918 is connected to Vs+. However, the output voltage Vh 920 will bedriven to approximately 4*Vs+ if Vg 918 is connected, instead, to Vd910. A storage capacitor, though not shown, is generally providedsomewhere to store and smooth each of the voltages Va 904, Vc 908, Vd910, Ve 914, Vg 918 and Vh 920.

The circuit of FIG. 9 may also be configured to produce negativevoltages. Vh 920 may be connected to a positive source voltage Vs+,while Vg 918 and Ve 914 are connected to a reference (ground) voltage,to produce −(Vs+) at Vf 916. Vc 908 and Va 904 may be connected to Vf916. An output Vb 906 will be approximately −2*Vs+ if Vd 910 isconnected to ground. If, instead, Vd 910 is connected to Vs+, then Vd910 will be driven to approximately −3*Vs+. Capacitive storage, notshown, is assumed for each supply and intermediate voltage.

In FIG. 9, all of the p-switches 800 and all of the n-switches 830 maybe controlled by a single phase of a clock, such as CLK 524 of FIG. 5.Alternatively, a single phase of a first clock may control the switchesconnected to the fly capacitor 912 (i.e., the switches of a charge pumpsection 922), while a single phase of a different second clock maycontrol the switches connected to the fly capacitor 902 (i.e., theswitches of a charge pump section 924). If the clocks for charge pumpsections 922 and 924 are inverted from one another, then the charge pumpsections may be disposed in parallel to create a push-pull charge pump.That is, Va 904 may be connected to Ve 914, Vb 906 to Vf 916, Vc 908 toVg 918, and Vd 910 to Vh 920. Such a push-pull charge pump may, forexample, provide lower ripple voltage and higher output currentcapabilities.

Additionally, in circuits otherwise comporting with FIG. 9, one or morepassive switches (such as switches 704 or 706 in FIG. 7) may replaceswitches in FIG. 9 that are shown (in FIG. 8) to be controlled by acharge pump clock output signal coupled to a control node of the switch.For example, the p-switch connected to Vg 918 and/or the n-switchconnected to Vf 916 may be replaced by a passive switch, or by a diode.Similar substitutions may be made for the switches connected to Vc 908and/or to Vb 906.

Moreover, if Vd 910 and Va 904 are equivalent to Vin+ 402 and commonreference 404 of FIG. 5, then the left side of the transfer capacitor902 may be coupled directly to a charge pump clock output such as 524,which would omit the left-side switches of the pump section 924 in FIG.9. The left-side switches of section 922 of FIG. 9 may be omitted underanalogous circumstances, with the left side of the transfer capacitor912 coupled directly to a charge pump clock output instead. In such aconfiguration, transfer capacitor charging current and dischargingcurrent may be limited not only by a drive signal (such as that oncapacitor 522 in FIG. 5) that has a limited dv/dt, but further oralternately by a current limiting device such as 512 or 518.

FIG. 10 is a simplified schematic diagram that illustrates generation ofarbitrary voltages by means of charge pump techniques. A source voltageVs+ 1002 may be used to derive an intermediate voltage, for example bymeans of resistors 1004 and 1006. Such intermediate voltage mayoptionally be buffered, if significant current output is desired, bymeans of buffer amplifier 1008, or may at least be connected to groundvia a decoupling capacitor (not shown), to produce a projection voltage1010 (with respect to common, or ground). The projection voltage 1010may be regulated, for example by means of feedback (not shown) appliedto amplifier 1008. P-switches 800 and n-switches 830 may be employed inconjunction with a fly capacitor 1012, as shown. Vb may be an availablevoltage source (e.g., Vs+). Operation of the circuit will establish Va1014 at a voltage approximately equal to that of Vb 1017 plus theprojection voltage 1010. The skilled person will readily see thatrearrangement of the circuit will permit generation of arbitrarynegative voltages with equal simplicity. Larger voltages may be obtainedby stacking charge pump circuits, in a manner as described with respectto FIG. 9. Thus, arbitrary positive or negative voltages may begenerated by means of charge pump circuits. All such charge pumpcircuits may be controlled by a single phase of a clock. Of course, theswitches associated with different fly capacitors may alternatively usedifferent clocks for control. Active switches may also be replaced withpassive switches, for example in a manner as described with respect toFIG. 7 or FIG. 9.

Innumerable alternative embodiments of the charge pumps described aboveare possible. Any clock generator may be employed. It is desirable thatthe clock generator avoid introducing high di/dt noise on the supply, ascan be achieved with many analog designs (for example, as shown in FIG.5) and some digital designs. It is desirable that dv/dt of the clockdrive signals be limited, both to help limit control node signalinjection, and also to help limit (in conjunction with thetransconductance of the relevant switch) the resulting di/dt of chargingand discharging currents. It may be desirable to capacitively couple acharge pump clock to a control node of a transfer capacitor switch, forexample in order to simplify control of control-node voltage levels. Itmay be desirable to utilize a single-phase clock output to control many,or all, of the transfer capacitor active switches within a charge pump,to minimize circuit complexity. Any of these desirable features may beeffected independently, or in combination with any of the otherdesirable features, or in combination with any other feature describedherein. Thus, the skilled person may apply aspects of the method andapparatus described herein to a staggering variety of charge pumpconfigurations.

CONCLUSION

The foregoing description illustrates exemplary implementations, andnovel features, of aspects of a charge pump method and apparatus forgenerating new voltages in circuits. The skilled person will understandthat various omissions, substitutions, and changes in the form anddetails of the methods and apparatus illustrated may be made withoutdeparting from the scope of the invention. Numerous alternativeimplementations have been described, but it is impractical to list allembodiments explicitly. As such, each practical combination of apparatusand method alternatives that are set forth above or shown in theattached figures, and each practical combination of equivalents of suchapparatus and method alternatives, constitutes a distinct alternativeembodiment of the subject apparatus or methods. For example, a chargepump having a plural transfer capacitor (or ‘stacked’) architecture asshown in FIG. 9, having at least one passive switch as shown in FIG. 7,and having a settable voltage source as shown in FIG. 10, constitutes anembodiment of the subject apparatus, as do all other such practicalcombinations of subfeatures. Therefore, the scope of the presentedinvention should be determined only by reference to the appended claims,and is not to be limited by features illustrated in the foregoingdescription except insofar as such limitation is recited in an appendedclaim.

All variations coming within the meaning and range of equivalency of thevarious claim elements are embraced within the scope of thecorresponding claim. Each claim set forth below is intended to encompassany system or method that differs only insubstantially from the literallanguage of such claim, as long as such system or method is not, infact, an embodiment of the prior art. To this end, each describedelement in each claim should be construed as broadly as possible, andmoreover should be understood to encompass any equivalent to suchelement insofar as possible without also encompassing the prior art.

1. Charge pump apparatus for generating an output voltage supply withina circuit, comprising: a) a transfer capacitor; b) a plurality oftransfer capacitor coupling switches, each switchable between aconducting state and a nonconducting state under control of at least onecharge pump clock output; and c) a charge pump clock generating circuitincluding a ring oscillator comprising an odd number of not more thanthree inverting driver sections cascaded sequentially in a ring suchthat each driver section has an output coupled to a next driver sectioninput, wherein a first driver section is next after a last driversection and one of the driver section outputs constitutes a particularcharge pump clock output controlling at least one of the transfercapacitor coupling switches, and wherein each driver section includes i)circuitry configured as an active current limit to limit a rate of riseof voltage at the driver section output, and ii) circuitry configured asan active current limit to limit a rate of fall of voltage at the driversection output; d) wherein the plurality of transfer capacitor couplingswitches are coupled to the transfer capacitor, and are controlled so asto couple the transfer capacitor to a voltage source during periodicfirst times, and to couple the transfer capacitor to the output voltagesupply during periodic second times that are not concurrent with thefirst times.
 2. The apparatus of claim 1, wherein the plurality oftransfer capacitor coupling switches are under control of the particularcharge pump clock output.
 3. The apparatus of claim 1, wherein theactive current limit circuitry of (c)(i) and (c)(ii) is furtherconfigured to limit source and sink currents, conducted by each driversection within the charge pump clock generating circuit, tosubstantially identical magnitudes.
 4. Charge pump apparatus within amonolithic integrated circuit for generating an output voltage supply,comprising: a) a transfer capacitor coupled alternately between sourceconnections and output connections; b) a plurality of active switches,each switchable between a conducting state and a nonconducting stateunder control of at least one charge pump clock output to couple charge,which is not substantially conducted by the charge pump clock output,from the source connections to the output connections; c) a charge pumpclock generating circuit including an active driver circuit configuredto both source current to and sink current from the charge pump clockoutput to cause a voltage waveform of the charge pump clock output to besubstantially sine-like due to i) circuitry configured to limit sourcecurrent provided by the active driver circuit to the charge pump clockoutput, and ii) circuitry configured to limit current sunk from thecharge pump clock output by the active driver circuit.
 5. The apparatusof claim 4, wherein the charge pump clock generating circuit c) furthercomprises a discrete capacitive element coupled to the charge pump clockoutput and configured to reduce voltage rates of change at the chargepump clock output.
 6. The apparatus of claim 4, wherein the charge pumpclock generating circuit includes a plurality of active driver circuitseach configured to both source and sink current with respect to acorresponding driver output node, and wherein the charge pump clockgenerating circuit includes circuitry to limit the current sourcecapacity of each of the active driver circuits and circuitry to limitthe current sink capacity of each of the active driver circuits withrespect to the corresponding driver output node.
 7. A method ofgenerating an output supply from a charge pump incorporated within amonolithic integrated circuit by transferring charge from a sourcevoltage to a transfer capacitor (“TC”) alternately with transferringcharge from the TC to the output supply, wherein a TC-coupling switch(“TCCS”) circuit is a switching circuit of the charge pump configured tocouple the TC to a supply under control of a charge pump clock, themethod comprising: a) coupling the TC to the output supply duringdischarge periods via a discharging TCCS circuit under control of afirst charge pump clock output; and b) actively limiting a rate ofvoltage change of the first charge pump clock output during bothpositive transitions and negative transitions such that a voltage of thefirst charge pump clock output is substantially sine-like.
 8. The methodof claim 7, further comprising c) coupling the TC to the source voltagevia a charging TCCS circuit, under control of a second charge pump clockoutput, during charge periods that nonoverlappingly alternate with thedischarge periods; and d) actively limiting a rate of voltage change ofboth positive and negative transitions of the second charge pump clockoutput.
 9. The method of claim 8, wherein the first charge pump clockoutput is the second charge pump clock output.
 10. The method of claim9, further comprising controlling all TCCS circuits by means of thefirst charge pump clock output.
 11. The method of claim 10, furthercomprising coupling the TC to a connection of the source voltage duringa charging period via the charge pump clock output.
 12. The method ofclaim 7, further comprising limiting a current drive capacity of thecharge pump clock output by means of a current limiting circuit.
 13. Amethod of generating an output supply by alternately transferring chargefrom a source voltage to a transfer capacitor (“TC”), and from the TC tothe output supply, the method comprising: a) coupling the TC to theoutput supply during discharge periods via a discharging switch circuitunder control of a first charge pump clock output; b) limiting sourcecurrent provided to each inverting driver output node of acurrent-starved ring oscillator having not more than three invertingdriver stages within a first charge pump clock generator circuit bymeans of a corresponding source current-limiting circuit; and c)limiting sink current drawn from each of the inverting driver outputnodes by means of a corresponding sink current-limiting circuit; whereinthe inverting driver output node of one of the not more than threeinverting driver stages of the first charge pump clock generator circuitis the first charge pump clock output.
 14. The method of claim 13,further comprising d) coupling the TC to the source voltage via acharging switch circuit, under control of a second charge pump clockoutput, during charge periods alternating nonconcurrently with thedischarge periods.
 15. The method of claim 13, further comprisingcoupling a capacitor to the driver output node of the first charge pumpclock generating circuit to limit voltage transition rates of the driveroutput node.
 16. The method of claim 13, further comprising coupling thefirst charge pump clock output to a control node of the dischargingswitch circuit and/or to a control node of a charging switch via acorresponding capacitive coupling circuit.
 17. The method of claim 13,further comprising: c) coupling a second TC to a second voltage sourcevia a second TC charging switch under control of the charge pump clockoutput; and d) coupling the second TC to a second output supply via asecond TC discharging switch under control of the charge pump clockoutput.
 18. The apparatus of claim 1, wherein the number of driversections of the ring oscillator is not less than three.